Driver circuit for switching converters, corresponding converter and method

ABSTRACT

A driver circuit includes an input node to receive an input signal for conversion at the output node of a converter, a driver node to provide to a switching power circuit stage in the converter a pulse-width modulated drive signal having an active time, first and second active time generation paths, and a selector circuit coupled to the first and second active time generation paths. The circuit is operable selectively in a first and a second operational mode wherein the driver node receives the pulse-width modulated drive signal having a first active time value generated in the first active time generation path, or a second active time value generated in the second active time generation path. The second active time generation path includes an active time generator network to provide a second active time value with the second active time value adaptively variable to match the first active time value.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to Italian Patent Application No. 102018000009321, filed on Oct. 10, 2018, which application is hereby incorporated herein by reference.

TECHNICAL FIELD

The description relates to switching converters.

One or more embodiments may be applied to systems with low energy consumption, a factor of strong topicality in the current electronics market.

BACKGROUND

Converters such as DC-DC converters comprising a Pulse Skipping Modulator (PSM) facilitate reducing switching losses at low (light) loads while being able to preserve the overall efficiency of the converter and maintain the synchronicity with a clock reference. On the other hand, pulse skipping operation may result in output voltage ripple related primarily to external components, supply voltage and losses, thus being difficult to control.

SUMMARY

Despite the extensive activity in the area, further improved solutions are desirable.

For instance, solutions are desirable which may be capable of optimizing output voltage ripple, and which may exhibit adequate immunity to noise, stability over a wide range of system variables and, possibly, a low impact on static consumption.

An object of one or more embodiments is to contribute in providing such an improved solution.

According to one or more embodiments, such an object can be achieved by means of a circuit having the features set forth in the claims that follow.

One or more embodiments may relate to a corresponding converter.

One or more embodiments may relate to a corresponding method.

The claims are an integral part of the technical teaching provided herein in respect of the embodiments.

One or more embodiments may provide an “event driven” solution having one or more of the following advantages:

virtually no static current consumption, which facilitates increasing efficiency at low loads;

the duration of T_(ON) (that is, the active time and thus the duty-cycle) of the pulse-width modulated signal can be modulated as a function of system losses, which facilitates cancelling inaccuracies in the control loop and making state passages between low-power and high-power smoother;

reduced impact in terms of silicon area occupancy, with cost savings;

suitability for use in both synchronous and asynchronous systems, with improved adaptability to a wide range of applications;

high (notionally complete) immunity to noise achieved via a digital approach, which is a factor of primary importance for a wide range of products and applications.

One or more embodiments may be applied, for instance, in devices including one or more step-down DC-DC converters, leading to a desired behavior of the output voltage or at the switching node of the regulator and its coils current.

One or more embodiments may include digital circuits able to modulate the T_(ON) intended for the power section.

For instance, one or more embodiments may be able to adapt the conduction time with a certain granularity (for instance, with a digital approach) in response to variations in the voltage supply and/or load current, with duration maintained essentially equal to the switching period which is observed in a high-power consumption mode.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more embodiments will now be described, by way of example only, with reference to the annexed figures, wherein:

FIG. 1 is a block diagram of a pulse width modulation (PWM) control loop in a converter;

FIG. 2 includes two diagrams indicated a) and b), respectively, exemplary of moving from high-power to low-power operation mode in an arrangement as shown in FIG. 1 in the absence of an integrator;

FIG. 3 is a diagram showing certain signal behavior useful in understanding embodiments;

FIG. 4 includes six diagrams indicated a) to f), respectively, exemplary of operation of embodiments;

FIGS. 5 and 6 are block and circuit diagrams exemplary of embodiments;

FIG. 7 is a circuit diagram of a possible implementation detail of embodiments;

FIG. 8 is a set of time diagrams of signal behavior related to FIG. 7;

FIGS. 9 and 10 each include four time diagrams, exemplary of possible operation of embodiments; and

FIG. 11 is a circuit diagram of a possible implementation detail of embodiments.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

In the ensuing description, one or more specific details are illustrated, aimed at providing an in-depth understanding of examples of embodiments of this description. The embodiments may be obtained without one or more of the specific details, or with other methods, components, materials, etc. In other cases, known structures, materials, or operations are not illustrated or described in detail so that certain aspects of embodiments will not be obscured.

Reference to “an embodiment” or “one embodiment” in the framework of the present description is intended to indicate that a particular configuration, structure, or characteristic described in relation to the embodiment is comprised in at least one embodiment. Hence, phrases such as “in an embodiment” or “in one embodiment” that may be present in one or more points of the present description do not necessarily refer to one and the same embodiment. Moreover, particular conformations, structures, or characteristics may be combined in any adequate way in one or more embodiments.

The references used herein are provided merely for convenience and hence do not define the extent of protection or the scope of the embodiments.

The block diagram of FIG. 1 is exemplary of a conventional DC-DC step-down converter including the following elements:

10: a loop comparator wherein an output voltage V_(OUT) from the converter is compared with a reference voltage V_(REF);

12: a T_(ON) generation circuit sensitive to the output from the loop comparator 10 and configured to provide a pulse-width modulated signal PWM with an active time T_(ON), a non-active time T_(OFF) and a duty cycle T_(ON)/(T_(ON)+T_(OFF));

14: a switching power stage (for instance a power MOSFET stage) driven by the signal PWM from the T_(ON) generation circuit 12 and supplied by an input (voltage) V_(IN);

16: a V_(IN) feed forward stage which applies the input signal V_(IN) the T_(ON) generation circuit 12;

18: a zero current (ZC) comparator active between the power stage 14 and the T_(ON) generation circuit 12; and

20: a low-pass output filter network, for instance a LC low-pass filter including inductive and capacitive components L_(OUT) and C_(OUT), which provides the output (voltage) signal V_(OUT) from the switching output V_(SW) of the power stage 14.

As noted, an arrangement as exemplified in FIG. 1 is conventional in the art, which makes it unnecessary to provide a more detailed description herein.

In an arrangement as exemplified in FIG. 1, the pulse-width modulated signal PWM generated at 12 (and primarily the active time or On-time T_(ON) thereof) is triggered via a feedback loop through the comparator 10 as soon as the difference V_(OUT)−V_(REF) is equal to zero and is a function of the input signal V_(IN) and V_(OUT) itself.

The arrangement of FIG. 1 is exemplary of a solution where, in order to save power dissipation at (ultra) low loads, an integrator along the control loop from the output V_(OUT) is replaced by a simple loop comparator (that is the comparator 10) which determines the need for charge to be possibly supplied on the output.

An arrangement as exemplified in FIG. 1 does not include (active) circuitry able to finely modulate the duration of the “on” time T_(ON) in the signal PWM. This is thus generally forced in open loop (via 16) with possible inaccuracies resulting in terms of output voltage ripple, current ripple in the coil (L_(OUT)), with the state passages (high-power/low-power) in the switching power stage 14 far from being smooth.

Heavy trimmings may be adopted in the T_(ON) generation circuits (in both 12 and 16) with the aim to mitigate the intrinsic inaccuracy related to low-power operation in the absence of an integrator along the control loop. This may in turn involve a notable waste in silicon area occupied.

Also, in those applications where appreciable variations in the supply voltage may be expected, a feed-forward loop may be included in order to track these variations in support of the T_(ON) generation circuits. This inevitably (further) weighs down the budget in terms of dissipated power and area occupancy on silicon.

An arrangement which facilitates making state passages smooth, with the ability of modulating the conduction time of a buck converter also in a low-power mode and reducing the output ripple is disclosed in US 2015/0137776 A1.

While satisfactory under various aspects, it is noted that such an arrangement may:

involve an error with a dedicated compensation circuit along the loop;

exhibit current consumption which makes it hardly suitable for certain low-power applications, such as applications involving a “deep sleep mode” for client SSD devices. Client solid state drive (c-SSD) is a designation oftentimes adopted to distinguish between solid state drives for the consumer market and solid state drives for the professional market;

have a high area occupancy with a negative impact on cost;

be suited to operate only within a pulse skip modulator (PSM), where a reference clock is present to dictate the switching times, which is hardly compatible with a pulse frequency modulation (PFM) approach,

exhibit noise sensitivity due to the (totally) analog approach.

For instance, when the load current falls to values of the order of a mA (1 milliampere=10⁻³ A) or of the hundreds of μA (1 microampere=10⁻⁶ A), a few μA of static current can have a significant impact on the overall system efficiency of, for instance, a so-called Buck regulator in PSM mode.

More to the point, it is noted that:

in a high consumption mode, the presence of an integrator may facilitate cancellation of the inaccuracies present in the system;

in a low consumption mode, if an open loop approach is chosen in generating T_(ON), these inaccuracies directly affect the output voltage and more deeply its ripple.

The state passage between the two modes (high consumption/power and low consumption/power) may a critical aspect for the application.

For instance, FIG. 2 is schematically exemplary of a possible behavior of the current I_(COIL) through L_(OUT)—upper diagram of portion a)—through L_(OUT) and the voltage V_(OUT)—lower diagram of portion b)—when transitioning between high-power operation (HP, left-hand side) and low-power operation (LP, right-hand side).

One or more embodiments may address the issues discussed previously by resorting to an “event driven” digital approach in generating the On-time (active time T_(ON) of the pulse-width modulated signal PWM), for instance in low-power conditions.

Such an event-driven solution is found to exhibit notionally zero static consumption due to consumption confined to a commanded On-time action performed.

One or more embodiments may involve a ring oscillator and a counter which facilitate modulating the On-time duration.

A corresponding step size can be selected small enough—for instance, 0.7 ns (1 ns=10⁻⁹ s)—to avoid steady-state subharmonics, with a corresponding digital word (for instance 12 bits) used to determine the On-time duration which is compatible with system requirements.

FIG. 3 is exemplary of such an approach and shows (with respect to a common time scale t) possible options in generating such a signal DTON related to T_(ON) TRIGGER and T_(ON) HIGH-POWER pulses.

As exemplified in FIG. 3, with a converter (regulator) in a high-power mode, the digital signal DTON can be increased (if the pulse on signal DTON is detected to be shorter than T_(ON) HIGH-POWER), decreased (if the pulse on signal DTON is detected to be greater than T_(ON) HIGH-POWER) or maintained unaltered (if the pulse on signal DTON has the same duration of those on signal T_(ON) HIGH-POWER) at each conduction time by comparing the effective On-time at the power stage 14 (generally dictated by an integrator along the control loop) with a value for T_(ON) generated digitally by the circuit. This facilitates causing the first T_(ON) performed when transitioning to low-power mode to have complete consistency and continuity with the last one performed in the (preceding) high-power mode, so that the transitions between the states (high-power to low-power) may be smooth and exempt from ripple jumps.

With the converter is a low-power mode, the digital word which dictates the duration of the pulse on signal DTON can be modulated in order to produce current pulses lying exactly an ideal high-power switching period (T_(SW)).

Thus:

if a zero current (ZC) event (in the power stage 14) during the Off-time (or T_(OFF)) is detected too late or too early, the following digital On-time will be reduced or increased by a single step,

if the zero current rising edge occurs at the end of the ideal T_(SW), the following digital On-time and the associated digital word remain unchanged.

As an example, in the presence of a reduction in supply (for instance, a drop in the input voltage V_(IN)) the system reacts by increasing the On-time by a single step at each pulse in order to restore the condition where the conduction time is ideally equal to a single T_(SW).

The information used to discriminate whether updating the digital word is desirable or not may be provided by the position of a zero current event (ZC) in the ideal high-power T_(SW): more to the point, a corresponding routine can be devised which stops as a result of such an event triggered in the last portion of the period (hence the desirability of a high frequency clock—12 MHz, for instance—capable of finely partitioning the associated time window).

A corresponding example is depicted in FIG. 4, where the uppermost diagram—labeled a)—is exemplary of a decrease of the input voltage V_(IN) to the converter from V_(IN,1) to V_(IN,2) with V_(IN,2)<V_(IN,1).

The diagrams labeled b) to f) show—against a common time scale t—possible behaviors of certain signals coming along with the decrease of the input voltage V_(IN) from V_(IN,1) to V_(IN,2):

b) T_(ON) TRIGGER

c) HIGH-POWER T_(SW)

d) the current I_(COIL) in the coil L_(OUT)

e) the output of the zero current detector (Zero Current)

f) a corresponding clock signal CLK (for instance at 12 MHz).

The portions of the diagrams c) to f) reproduced on an enlarged scale in the dashed line frames indicated as 4A and 4B highlight the behavior of the routine just described in keeping the coil current I_(COIL) (which may have the sawtooth-like behavior exemplified between 4A and 4B) within a desired time frame as exemplified by HIGH-POWER T_(SW) irrespective of the reduction of the input voltage V_(IN) from V_(IN,1) to V_(IN,2).

One or more embodiments may thus be regarded as exemplary of solutions that implements a sort of slow supply feed-forward (at no cost in terms of dissipated power).

FIG. 5 is a block diagram exemplary of embodiments.

For brevity and simplicity, parts or elements like part or elements already discussed in connection with FIG. 1 are indicated in FIG. 5 with like reference symbols, thus making it unnecessary to repeat a detailed description.

Briefly, the block diagram of FIG. 5 can be regarded as exemplary of a converter circuit as introduced in FIG. 1 wherein the T_(ON) generation circuit block 12 (and the V_(IN) feed forward circuit block 16) are replaced with circuitry designated 120 as a whole.

Circuitry 120 as discussed in the following may be configured to generate the pulse-width modulated signal PWM fed to the power circuits stage 14 according to the principles introduced in the foregoing in connection with FIGS. 3 and 4.

The circuitry in the block diagram of FIG. 5 can be regarded as including a high-power loop and a low-power loop intended to provide respective values for the “on” signal T_(ON), namely T_(ON) HIGH-POWER and DTON, to a selector circuit 122. The circuit 122 may thus forward to the power circuit stage 14 (as a PWM drive signals therefor) either one of T_(ON) HIGH-POWER or DTON as a function of a state signal HIGH/LOW-POWER supplied to the selector circuit 122 by a (main) logic circuit 22.

In one or more embodiments, the logic circuit 22 (a microcontroller, for instance) may be configured—in a manner known to those of skill in the art—to provide:

a first (reference) clock signal CLK_REF to a high-power T_(ON) generator 124 in the high-power loop which is sensitive to the input voltage V_(IN) and provides to the selector circuit 122 the T_(ON) HIGH-POWER signal;

a second (for instance 12 MHz) clock signal CLK_12MHZ and an (ideal) T_(SW) signal to an “event driven” circuit block 126 in the low-power loop which is configured to provide to the selector circuit 122 the DTON signal as further discussed in the following;

the HIGH/LOW-POWER signal indicative of the converter circuit being in a high-power or low-power operational state with this signal supplied also to the event driven circuit block 126 in addition to the selector circuit 122.

As exemplified in FIG. 5, the event driven circuit block 126 is also sensitive to:

a loop compensation signal LOOP_COMP from the loop comparator 10 where the output voltage V_(OUT) is compared with the reference voltage V_(REF);

the output ZERO_CURRENT from the zero current comparator 18;

the T_(ON) HIGH-POWER signal from the high-power T_(ON) generator 124 (which may result in the circuit block 126 being sensitive to the reference clock signal CLK_REF).

In a circuit as exemplified in FIG. 5 the high-power T_(ON) generator 124 in the high-power loop may also be sensitive to the output signal from a differential circuit 24 in a (negative) feedback configuration, wherein the differential circuit 24 receives at the non-inverting input the reference voltage V_(REF) and is coupled at the inverting input to a compensation network 26 sensitive to the input to the high-power T_(ON) generator 124.

The blocks 24 and 26 facilitate increasing the DC gain of the high-power loop compensating the inaccuracies and guaranteeing at the same time the stability as soon as L_(OUT) and C_(OUT) vary.

The circuit diagram of FIG. 6 is a low-level representation of a possible implementation of the circuit block 126 intended to generate the signal DTON, with the same signal designations already introduced in the foregoing reproduced for simplicity.

It will be otherwise appreciated that both FIGS. 5 and 6 contemplate the possible presence of a reference clock, namely CLK_REF. This may be used also in the low-power mode, as an input for the circuit block 126 (for instance via the circuit block 124).

In fact, in addition to a purely asynchronous low-power control, one or more embodiments may be suited for use also in a (totally) synchronous system.

Both these options will be discussed in the following.

In the circuit diagram of FIG. 6, reference 1260 denotes a T_(ON) request management block sensitive to the signal LOOP_COMP and configured to be activated (only) in the low-power mode with the role of transforming T_(ON) requests from the loop comparator 10 into small 0→1→0 pulses in order to start an internal oscillator 1262. A ring oscillator as discussed in connection with FIG. 7 may be exemplary of such an oscillator.

Depending on the type of the control loop (for instance: asynchronous or synchronous, continuous-conduction mode—CCM allowed in addition to discontinuous-conduction mode DCM, . . . ) the sub-block 1260 may also be made sensitive to the ZERO_CURRENT and CLK_REF signals.

For instance, in the case of a synchronous system forced to operate in discontinuous-conduction mode (DCM) during low-power mode operation, all the three signals LOOP_COMP, ZERO_CURRENT and CLK_REF may be used to start an On-time. Conversely, in an asynchronous system free to evolve in Continuous-Conduction Mode (CCM) just the LOOP_COMP signal may be used to start a charging phase.

Once a T_(ON) trigger pulse has been produced the oscillator 1262 and an associated counter 1264 cascaded thereto exit their reset condition and an internal DTON signal, designated iDTON in FIG. 6, is set to be input to a digital T_(ON) word update and actuation circuit block 1266.

In one or more embodiments as exemplified in FIG. 6, the oscillator 1262 may have an intrinsic clock period equal to, for instance, (0.7 ns×4×2)=5.6 ns (1 ns=1.10⁻⁹ s): this is of course a merely exemplary value.

In one or more embodiments, a clock pre-selector circuit block 1262 a may be provided at the output of the oscillator 1262.

The pre-selector circuit block 1262 a may be configured to use, for instance, 3 of the 12 bits of the T_(ON) digital word (that is a time resolution equal to 0.7 ns as shown in FIG. 7). This approach facilitates finely modulating the digital T_(ON) duration saving area and energy.

In an embodiment as exemplified in FIG. 7 (right hand side) the clock pre-selector circuit block 1262 a may include a multiplexer coupled to the oscillator ring and controlled by a first selection signal DTON_SEL[1:0] as well as an EX-OR gate having inputs coupled to the output from the multiplexer and to a second selection signal DTON_SEL[2] as well as providing a (selected) clock signal CLK to the counter 1264.

The selection signals DTON_SEL[1:0] and DTON_SEL[2] can be provided (in a manner known per se) by the actuation logic 1266 which may also provide a corresponding signal DTON_SEL[11:3] to the counter 1264.

In one or more embodiments the counter 1264 following the pre-selector circuit block 1262 a may be a 9-bit counter, which reaches its overflow more or less later depending on the T_(ON) digital word status.

The overflow condition of the counter 1264 determines the falling edge of iDTON signal and the (automatic) reset of the state machine embodied in the circuit 126.

When in the low-power mode, the signal iDTON is managed (directly) by the actuation logic 1266 to generate the DTON signal intended to be applied (via the selector 122) to the power stage circuitry 14.

Conversely, in the high-power mode, the high duration of this signal is compared with the value for T_(ON) generated along the control loop (with possible presence of an integrator) with the purpose of aligning the “digital” On-time generated at 126 with the “analog” On-time generated at 124, for instance by increasing or reducing adequately (see also FIG. 3) the digital word in the circuit 126.

Fixing, for instance via a delay network designated 1266 a in FIG. 6, a certain delay (5.6 ns, in the instant exemplary case) between an UP signal path and a DOWN signal path for T_(ON) HIGH-POWER to the actuation logic 1266 may facilitate dispensing with steady-state jitter on the T_(ON) digital word.

In embodiments as exemplified herein, in high-power mode operation, the digital T_(ON) duration (DTON from 126) tracks the analog T_(ON) duration (T_(ON) HIGH-POWER from 124) provided by the main control loop.

In embodiments as exemplified herein, in low-power mode operation—in case an asynchronous control is applied—at each T_(ON) trigger pulse, a time window is opened having a duration notionally equal to an “ideal” high-power switching period T_(SW). In addition, also an internal clock (12 MHz in the example considered herein) will be running during the same window to monitor the instant at which a rising edge occurs in the ZERO_CURRENT signal.

In an example as considered herein, three different cases are possible:

the ZERO_CURRENT rising edge occurs after the T_(SW) mask ending; the T_(ON) digital word is decreased by one step (see also the second time diagram in FIG. 3);

the ZERO_CURRENT rising edge occurs immediately before the T_(SW) mask ending, during the last 12 MHz clock period included within the T_(SW) mask; the T_(ON) digital word remains unchanged (see also the last time diagram in FIG. 3);

the ZERO_CURRENT rising edge occurs more than a 12 MHz clock period before the T_(SW) mask ending; the T_(ON) digital word is increased by one step (see also the next-to-last time diagram in FIG. 3).

In embodiments as exemplified herein, in low-power mode operation—in case a synchronous control is applied—the notional value for high-power T_(SW) is simply replaced by the (time) distance between two subsequent rising edges of the reference clock CLK_REF and the digital word update procedure (in low-power mode) will take into account also the condition of the converter (regulator) during the immediately preceding switching period with respect to the one where the zero current position is checked in the signal ZERO_CURRENT.

Considering a “good” pulse for the measurement may be facilitated by referring to On-time pulses triggered with the regulator output in high impedance condition.

For instance, all the pulses but the first one in a train of contiguous pulses can be discarded (it is noted that understanding simply if the pulses fall exactly within a certain switching period may be hardly possible).

In that respect, one may refer to the time diagrams a) to d) in FIG. 9 which (as in the case of FIG. 10) show, with reference to a common time scale t, a possible time behavior of the following signals:

CLK_12 MHz

CLK_REF

I_(COIL) (the current in the coil L_(OUT))

ZERO_CURRENT.

FIG. 9 is exemplary of a case where the first current pulse starts with the regulator (converter) output in high impedance with the second pulse, starting on the trailing edge of the first pulse.

This causes the current to increase starting from a point above zero. At the end of the switching period, the current is well above zero but it is not possible to define if the On-time duration was too high.

For that reason, in one or more embodiments, the second pulse will be discarded from the detection of the TON duration.

Conversely, FIG. 10 is exemplary of a case where the second pulse starts from a high impedance condition and thus also the second pulse is considered for the On-time duration measurement.

The circuit diagram of FIG. 11 is exemplary of a possible implementation of a circuit 1266 b which may be included in the circuit block 1266 in order to generate the signal DTON starting from the “internal” signal iDTON.

This facilitates complying with minimum On-time and Off-time values as desirable for adequate operation of the power circuit stage 14.

As exemplified in FIG. 11, signal iDTON can be applied to one input of an OR gate O₁ whose other input receives the output signal DTON through the cascaded arrangement of a delay element TD (configured to provide and asymmetrical rise/fall minimum T_(ON)/T_(OFF) delay) and an inverter I₁. The output from the OR gate O₁ is passed to a first input A of a (combinatorial) logic network C whose other input, designated B, is provided by the output for man AND gate A₁ which receives as inputs the signal T_(ON) TRIGGER (input to the oscillator 1262 as shown in FIG. 7) and the signal DTON delayed at TD and inverter at I₁.

A possible truth table (output Z v. inputs A and B) for the network C may be as follows:

A B Z 0 0 0 0 1 Z⁻¹ 1 0 Z⁻¹ 1 1 1

where Z⁻¹ denotes the previous value assumed by output Z.

Embodiments of an asynchronous DC-DC buck regulator as exemplified herein were found to have a current consumption at 85° C. (static plus dynamic) in the order of 320 nA (1 nA=10⁻⁹ A) in the following operating scenario:

deep low-power mode (I_(LOAD)=1 mA) V _(IN)=3·3V,V _(OUT)=1V

high-power T_(SW)=2 MHz

external inductor (L_(OUT)) 1 μH (1 μH=10⁻⁶ H), external capacitance (C_(OUT)) 44 μF (1 μF=10⁻⁶ F).

It is noted that consumptions figure indicated is more than one order lower compared to an “aggressive” analog solutions.

Area estimations for a totally synchronous system are in range of 16,900 μm² (1 μm=10⁻⁶ m).

In one or more embodiments, a circuit may comprise:

an input node configured to receive an input signal (for instance, V_(IN)) for conversion in a converter having an output node (for instance, V_(OUT));

a driver node configured to provide to a switching power circuit stage (for instance, 14) in the converter a pulse-width modulated drive signal (PWM) having an active time (for instance, T_(ON)),

first (for instance, 124) and second (for instance, 126) active time generation paths,

a selector circuit block (for instance, 122) coupled to the first and second active time generation paths as well as to the driver node, the selector circuit block configured to transfer to the driver node:

i) in a first operational mode of the circuit, a pulse-width modulated drive signal having a first active time value (for instance, T_(ON) HIGH-POWER) generated in the first active time generation path, or

ii) in a second operational mode of the circuit, a pulse-width modulated drive signal having a second active time value (for instance, DTON) generated in the second active time generation path,

wherein:

the first active time generation path comprises an active time generator circuit configured to provide said first active time value,

the second active time generation path comprises an active time generator network configured to provide a digitally-generated second active time value with said second active time value adaptively variable to match said first active time value as a result of the circuit transitioning between the first operational mode and the second operational mode.

In one or more embodiments, the active time generator network in the second active time generation path may be configured to provide step-wise variations of said second active time value, optionally by providing said second active time value as a digital word.

In one or more embodiments, the active time generator network in the second active time generation path may be configured (for instance, 1266) to increase (for instance, if DTON<T_(ON) HIGH-POWER), decrease (for instance, if DTON>T_(ON) HIGH-POWER) or maintain (for instance, DTON=T_(ON) HIGH-POWER) said digitally-generated second active time value as a result of the circuit transitioning from the first operational mode to the second operational mode.

In one or more embodiments, the active time generator network in the second active time generation path may comprise:

a generator circuit stage (for instance, 1260, 1262 a, 1264) configured to provide a respective digital value (for instance, iDTON) for said active time (for instance, T_(ON)) of the pulse-width modulated drive signal (PWM),

a comparator circuit (for instance, 1266) sensitive (for instance, 18, ZERO_CURRENT) to said respective digital value (iDTON) and a current effective value of said active time at said power circuit stage, the comparator circuit configured to perform a comparison of said respective digital value and said current effective active time value,

wherein the active time generator network in the second active time generation path is configured to increase (for instance, if DTON<T_(ON) HIGH-POWER), decrease (for instance, if DTON>T_(ON) HIGH-POWER) or maintain (for instance, DTON=T_(ON) HIGH-POWER), respectively, said digitally-generated second active time value as a function of the result of the comparison performed in said comparator circuit.

In one or more embodiments, the comparator circuit may be sensitive (for instance, 18, ZERO_CURRENT) to zero current events at said power circuit stage, wherein the active time generator network in the second active time generation path may be configured to:

maintain said digitally-generated second active time value as a result of zero current events occurring at said power circuit stage at controlled switching times (for instance, T_(SW)),

increase (for instance, if DTON<T_(ON) HIGH-POWER) or decrease (for instance, if DTON>T_(ON) HIGH-POWER) said digitally-generated second active time value as a result of zero current events occurring earlier or later, respectively, than said controlled switching times).

In one or more embodiments, the active time generator network in the second active time generation path may be configured to modulate said digitally-generated second active time value to match a reference switching period as a result of the circuit transitioning from the second operational mode to the first operational mode.

In one or more embodiments, a converter, may comprise:

a circuit according to one or more embodiments comprising said input node configured to receive said input signal (for instance, V_(IN)) for conversion in the converter having an output node (for instance, V_(OUT));

a switching power circuit stage (for instance, 14) coupled to the driver node of said circuit to receive therefrom said pulse-width modulated drive signal (PWM) having an active time (for instance, T_(ON)),

a converter output network, optionally an LC low-pass network (for instance, L_(OUT), C_(OUT)), comprising said output node, the converter output network driven by said power circuit stage.

One or more embodiments may comprise:

a reference node configured to receive a reference signal (for instance, V_(REF)),

a feedback line coupled to the output node from the converter to provide a feedback signal indicative of the signal at said output node,

and wherein said first and second active time generation paths may comprise differential input circuitry coupled to said reference node and said feedback line and configured to perform a comparison of said reference signal with said feedback signal and drive said first and second active time generation paths as function of the outcome of said comparison.

One or more embodiments may comprise an operation mode selection logic (for instance, 22), the operation mode selection logic configured to selectively switch operation of the converter between said first operational modes of the circuit and said second operational mode of the circuit.

In one or more embodiments, a method of operating the circuit or the converter of one or more embodiments may comprise selectively switching operation of said circuit between said first operational mode and said second operational mode.

Without prejudice to the underlying principles, the details and embodiments may vary, even significantly, with respect to what has been described by way of example only, without departing from the extent of protection.

The extent of protection is defined by the annexed claims. 

What is claimed is:
 1. A circuit, comprising: an input node configured to receive an input signal for conversion in a converter having an output node; a driver node configured to provide to a switching power circuit stage in the converter a pulse-width modulated drive signal having an active time; first and second active time generation paths; and a selector circuit having first and second inputs directly coupled to the first and second active time generation paths, and an output directly coupled to the driver node, the selector circuit configured to transfer to the driver node: in a first operational mode of the circuit, a pulse-width modulated drive signal having a first active time value generated in the first active time generation path, or in a second operational mode of the circuit, a pulse-width modulated drive signal having a second active time value generated in the second active time generation path, wherein: the first active time generation path comprises a first active time generator circuit configured to provide said first active time value, the second active time generation path comprises a second active time generator circuit configured to provide a digitally-generated second active time value with said second active time value that is adaptively variable to match said first active time value when the circuit transitions between the first operational mode and the second operational mode, the first active time generation path is configured for enablement in the first operational mode only, and the second active time generation path is configured for enablement in both of the first and second operational modes, and an output of the first active time generator circuit is coupled to an input of the second active time generator circuit.
 2. The circuit of claim 1, wherein the second active time generator circuit in the second active time generation path is configured to provide step-wise variations of said second active time value by providing said second active time value as a digital word.
 3. The circuit of claim 1, wherein the second active time generator circuit in the second active time generation path is configured to increase, decrease or maintain said digitally-generated second active time value when the circuit transitions from the first operational mode to the second operational mode.
 4. The circuit of claim 3, wherein the second active time generator circuit in the second active time generation path comprises: a generator circuit stage configured to provide a respective digital value for said active time of the pulse-width modulated drive signal; and a comparator circuit sensitive to said respective digital value and a current effective value of said active time at said switching power circuit stage, the comparator circuit configured to perform a comparison of said respective digital value and said current effective active time value.
 5. The circuit of claim 4, wherein the comparator circuit is sensitive to zero current events at said switching power circuit stage, wherein the second active time generator circuit in the second active time generation path is configured to: maintain said digitally-generated second active time value as a result of zero current events occurring at said switching power circuit stage at controlled switching times, and increase or decrease said digitally-generated second active time value as a result of zero current events occurring earlier or later, respectively, than said controlled switching times.
 6. The circuit of claim 1, wherein the second active time generator circuit in the second active time generation path is configured to modulate said digitally-generated second active time value to match a reference switching period when the circuit transitions from the second operational mode to the first operational mode.
 7. The circuit of claim 1, wherein the converter further comprises: a switching power circuit stage coupled to the driver node of said circuit to receive therefrom said pulse-width modulated drive signal having an active time; and a converter output network comprising an LC low-pass network and said output node, the converter output network being driven by said switching power circuit stage.
 8. The circuit of claim 7, wherein the converter further comprises: a reference node configured to receive a reference signal; and a feedback line coupled to the output node from the converter to provide a feedback signal indicative of the signal at said output node, wherein said first and second active time generation paths comprise differential input circuitry coupled to said reference node and said feedback line and configured to perform a comparison of said reference signal with said feedback signal and drive said first and second active time generation paths as function of the outcome of said comparison.
 9. The circuit of claim 7, wherein the converter further comprises an operation mode selection logic, the operation mode selection logic configured to selectively switch operation of the converter between said first operational mode of the circuit and said second operational mode of the circuit.
 10. The circuit of claim 1, wherein the circuit is configured for selectively switching operation between said first operational mode and said second operational mode.
 11. A method, comprising: receiving an input signal at an input node for conversion in a converter having an output node; providing a pulse-width modulated drive signal having an active time to a switching power circuit stage in the converter; providing first and second active time generation paths; in a first operational mode, selecting a pulse-width modulated drive signal having a first active time value generated in the first active time generation path; and in a second operational mode, selecting a pulse-width modulated drive signal having a second active time value generated in the second active time generation path, wherein the first active time generation path comprises a first active time generator circuit to provide said first active time value, wherein the second active time generation path comprises a second active time generator circuit to provide a digitally-generated second active time value with said second active time value that is adaptively variable to match said first active time value when the method transitions between the first operational mode and the second operational mode, wherein the first active time generation path is enabled in the first operational mode only, and the second active time generator path is enabled in both of the first and second operational modes, wherein an output of the first active time generator circuit is coupled to an input of the second active time generator circuit and directly coupled to an input of a selector circuit, and wherein an output of the selector circuit is directly coupled to the switching power circuit stage in the converter.
 12. The method of claim 11, wherein the second active time generator circuit in the second active time generation path provides step-wise variations of said second active time value by providing said second active time value as a digital word.
 13. The method of claim 11, wherein the second active time generator circuit in the second active time generation path increases, decreases or maintains said digitally-generated second active time value when the method transitions from the first operational mode to the second operational mode.
 14. The method of claim 13, wherein the second active time generator circuit in the second active time generation path provides a respective digital value for said active time of the pulse-width modulated drive signal, and performs a comparison of said respective digital value and a current effective active time value.
 15. The method of claim 14, wherein the second active time generator circuit in the second active time generation path maintains said digitally-generated second active time value as a result of zero current events occurring at said switching power circuit stage at controlled switching times, and increases or decreases said digitally-generated second active time value as a result of zero current events occurring earlier or later, respectively, than said controlled switching times.
 16. The method of claim 11, wherein the second active time generator circuit in the second active time generation path modulates said digitally-generated second active time value to match a reference switching period when the method transitions from the second operational mode to the first operational mode.
 17. The method of claim 11, further comprising low-pass filtering said output node.
 18. The method of claim 17, wherein said first and second active time generation paths perform a comparison of a reference signal with a feedback signal representative of an output signal at the output node and drive said first and second active time generation paths as function of the outcome of said comparison.
 19. The method of claim 11, further comprising selectively switching between said first operational mode and said second operational mode.
 20. A method, comprising: in a first operational mode, selecting a pulse-width modulated drive signal having a first active time value generated in a first active time generation path, wherein the first active time generation path comprises a first active time generator circuit to provide said first active time value; in a second operational mode, selecting a pulse-width modulated drive signal having a second active time value generated in a second active time generation path, wherein the second active time generation path comprises a second active time generator circuit to provide a digitally-generated second active time value with said second active time value being adaptively variable to match said first active time value when the method transitions between the first operational mode and the second operational mode; enabling the first active time generation path in the first operational mode only, and enabling the second active time generator path in both of the first and second operational modes; and coupling an output of the first active time generator circuit to an input of the second active time generator circuit; and using a selector circuit for selecting the first and second operational modes, wherein an output of the first active time generator circuit is directly coupled to a first input of the selector circuit, and an output of the second active time generator circuit is directly coupled to a second input of the selector circuit. 